Voltage References and Voltage (part 1)

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1. Introduction

Almost all electronic systems utilize a regulated power supply as an essential requirement. Most systems need a precision voltage reference as well. In the past, the task of voltage regulation was tediously accomplished with discrete devices.

Today, with integrated circuit voltage references and regulators, this task has been significantly simplified. Not only can an extremely high precision be obtained, but also an extremely high degree of temperature stability.

The performance of today's electronic devices such as microprocessors, test and measuring instruments, and sophisticated portable and handheld equipment is directly related to the quality of the supply voltage. This results in the need for tight regulation, low noise, and excellent transient response. The designer now has a wide choice of fixed, adjustable, and tracking voltage regulators, with many also incorporating built-in protection features.

One of the fastest growing markets in the world of power regulation is for switching regulators. These offer designers several important advantages over linear regulators, the most significant being size and efficiency. In addition, the ability to perform step-up, step-down, or voltage inverting functions is an attractive feature.

The old linear regulator is not totally out of business. The proliferation of battery-powered equipment in recent years has accelerated the development and usage of low-dropout (LDO) voltage regulators. Compared to a standard linear regulator, the LDO regulator using PNP transistors can maintain its output in regulation with a much lower voltage across it. While the NPN transistor requires about 2 V of headroom voltage to regulate, the LDO typically will work with less than 500 mV of input-to-output voltage differential. This reduced input voltage requirement is advantageous in battery-powered systems, since it translates directly into fewer battery cells (Simpson, 1996). In low-dropout applications, the efficiency advantage of switching regulators no longer is as great.

A linear regulator design on the other hand offers several desirable features, such as low output noise and wide bandwidth, resulting in excellent transient response.

This Section describes the basics of voltage references, linear and switching regulators, and continues to discuss the state-of-the-art components available, the advantages and disadvantages of different types of devices, their application environments as well as the basics of regulator design using these components.

2. Voltage References

2.1 Voltage Reference Fundamentals

A wide variety of voltage references are available today. However, all base their performance on the action of either a zener diode or a bandgap cell. Additional circuitry is included to obtain good temperature stability. Although discrete zener diodes are available in voltage ratings as low as 1.8 V to as high as 200 V, with power handling capabilities in excess of 100 W, their tolerance and temperature characteristics are unsuitable for many applications. Therefore, discrete zener diode-based references have additional circuitry to improve performance.

The most popular reference is probably the temperature-compensated zener diode, particularly, for voltages above 5V. The operation of a bandgap reference is based on specific characteristics of diodes operating at the same current but different current densities. Bandgap references are available with output voltage ratings of about 1.2 to 10 V. The principal advantage of these devices is their ability to provide stable low voltages, such as 1.2, 2.5, or 5 V. However, bandgap references of 5 V and higher tend to have more noise than equivalent zener-based references. This is because, in bandgap references, higher voltages are obtained by amplification of the 1.2 V bandgap voltage by an internal amplifier. Their temperature stability also is below that of zener-based references.

2.2 Types of Voltage References

2.2.1 Zener-Based Voltage References

Zener diodes are semiconductor PN junction diodes with controlled reverse- bias properties, which make them extremely useful as voltage references. The V-I characteristics of an ideal zener diode is shown in FIG. 1(a) and a simple regulator circuit based on it in FIG. 1 (b). The reverse characteristics show that, at the breakdown point, the knee voltage is independent of the diode current. This knee voltage or the zener voltage is controlled by the amount of doping applied in the manufacturing process. In the simple regulator circuit shown in FIG. 1(b), as long as the zener diode is in its regulating range, the load voltage VL remains constant and equal to the nominal zener voltage, even when the input voltage and the load resistance varies over a wide range. If the input voltage increases, the diode maintains a constant voltage across the load by absorbing the extra current and keeping the load current constant. If the load resistance decreases, the extra current required to keep the load voltage constant is facilitated by a decrease in the current drawn by the zener diode.


FIG. 1 Zener diode and voltage regulator (a) Typical zener characteristics (b) a simple zener diode voltage regulator


FIG. 2 Temperature characteristics of zener diodes: (a) Zener breakdown, (b) Avalanche breakdown

In the preceding simplified analysis, the temperature dependence of the zener voltage was not taken into account. The stability of the output with temperature is a prime requirement of a voltage reference. Not only does the zener voltage vary with temperature, its variation also depends on the type of breakdown that occurs.

A zener diode has two distinctly different breakdown mechanisms: zener breakdown and avalanche breakdown. The zener breakdown voltage decreases as the temperature increases, creating a negative temperature coefficient (TC). The avalanche breakdown voltage increases with temperature (positive TC). This is illustrated in FIG. 2. The zener effect and the avalanche effect dominate at low and high currents, respectively.

Although, theoretically, it’s possible to select the operating point of a zener diode so that the two temperature coefficients will cancel out each other, in practical IC zener-based voltage references, a conventional forward-biased diode is used in series with a zener operating in the avalanche mode. A forward-biased diode has a negative TC, and this cancels the positive TC of the zener diode.

A simple zener-based voltage reference IC is shown in FIG. 3. In this circuit, R4 provides the startup current for the diodes, thus setting the positive input of the opamp at V2. R3 sets the desired bias current for the diodes.

Manufacturers set the output voltage to a value different from that of V2 through the ratio R1 to R2. By trimming this resistor ratio, the output voltage can be set to the desired accuracy. Also, by trimming R3, the bias current can be optimized to a point where a minimum TC is obtained.

TC specifications as low as 1 ppm/°C are possible with zener-based voltage reference ICs (Pryce, 1990).


FIG. 3 A simple zener-based voltage reference IC


FIG. 4 The circuit diagram of a bandgap reference

2.2.2 Bandgap References

Similar to zener-based references, bandgap references also produce the sum of two voltages having opposite temperature coefficients. One voltage is the forward voltage of a conventional diode (the base-emitter junction of a transistor), which has a negative temperature coefficient. The other is the difference between the forward voltages of two diodes with the same current but operating at two current densities. A circuit diagram of a bandgap reference is shown in FIG. 4.

Transistors Q1 and Q2 are operating at the same current, but at different current densities. This is achieved by fabricating Q2 with a larger emitter area than Q1. Therefore, the base-emitter voltages of the two transistors are different.

This difference is dropped across R2.

Extrapolated to absolute 0, V_BE is equal to 1.205 V, the bandgap voltage of silicon, and has a predictable, negative temperature coefficient of -2 mV/°C. By adding a voltage to V_BE, which has a positive temperature coefficient, a bandgap reference, at least theoretically, can generate a constant voltage at any temperature.

The base-emitter voltage difference is given by:

Delta V_BE = In

where J1 and J2 are the current densities of transistors Q1 and Q2, respectively.

Since the sum of the two transistor currents flow through R1, the voltage across R1 can be expressed as:

V1 = 2 R2 Delta V_BE

Also, Using 1.2 and 1.3, V2 – V_BE --i- V1

V2=V_BE+2(R-~22) delta V_BE

Therefore, V2 is the sum of V_BE and the scaled Delta V_BE. Knapp (1998) shows that, if the emitter areas of the two transistors is eight, the temperature coefficients of V_BE and A V_BE cancel each other. The op amp raises the bandgap voltage V2 to a higher voltage at the output of the reference.

Bandgap references typically provide voltages ranging from 1.2 to 10 V. The advantage of bandgap references is their ability to provide voltages below 5 V. The greatest appeal of bandgap devices is the ability to function with operating currents from milliamps down to microamps.

IC bandgap references have additional features such as multiple calibrated voltages. Because most bandgap references are constructed in monolithic form, they are relatively inexpensive. However, their temperature coefficient is inferior to that of zener-based references. This is due to the second-order dependencies of A V_BE on temperature.

2.3 Quality Measures of Voltage References

An ideal voltage reference would have the exact specified voltage, and it would not vary with time, temperature, input voltage, or load conditions.

However, as it’s impossible to fabricate such ideal references, manufacturers provide specifications informing the user of the device's important quality parameters.

2.3.1 Output Voltage Error

This is the initial untrimmed accuracy of the reference at 25°C at a specified input voltage. This is specified in millivolts or a percentage. Some references provide pin connections for trimming their initial accuracy with an external potentiometer.

2.3.2 Temperature Coefficient

The temperature coefficient of a reference is its average change in output voltage as a function of temperature compared with its value at 25°C. This is specified in ppm/°C or mV/°C.

2.3.3 Line Regulation

This is the change in output voltage for a specified change in input voltage.

Usually specified in %/V or txV/V of input change, line regulation is a measure of the reference's ability to handle variations in supply voltage.

2.3.4 Load Regulation

This is the change in output voltage for a specified change in load current. Specified in uV/mA, %/mA, or ohms of DC output resistance, load regulation includes any self-heating effects due to changes in power dissipation with load current.

2.3.5 Long-Term Stability

This is the change in the output voltage of a reference as a function of time.

Specified in ppm/1000 hours at a specific temperature, the long-term stability is difficult to quantify. As a result, manufacturers usually provide only typical specifications, based on device data collected during the characterization process.

2.3.6 Noise

Although the preceding are the most important quality parameters of a voltage reference, noise is particularly of importance in certain applications such as A/D or D/A converters. In such applications, the noise from the reference should be less than 10% of the LSB value of the converter. Therefore, the higher the resolution of the converter, the lower should be the noise generated from the reference.

Noise depends on the operating current of the reference and generally is specified over a particular bandwidth and for a particular current. The specified bandwidths are 0.1-10 Hz (low-frequency noise) and 10 Hz-10 kHz (high-frequency noise).

2.4 Voltage Reference ICs

The levels of sophistication and pricing for voltage references range from simple and inexpensive to complex and costly. Devices are available for almost any conceivable application. Manufacturers of voltage references include National Semiconductor, Motorola, Analog Devices, Linear Technology, SGS-Thompson, Maxim Integrated Circuits, Texas Instruments, Precision Monolithic, and Silicon General.


TABLE 1 Illustrative Zener-Based References

2.4.1 Zener-Based References

Zener-based references usually are used in analog circuits that operate from 12-15 V supplies. Some zener-based voltage references are illustrated in TBL. 1.

A typical high-performance zener diode is the REF101 from Burr-Brown with a reference voltage of 10 V (Burr-Brown, 1989). The combination of its excellent parameters makes this device well suited for use with high-resolution A/D and D/A converters or as a precision calibrated voltage standard. This device has a very high accuracy of 0.005 V and a temperature drift of 1 ppm/°C. ~ Analog Devices offers a wide range of both zener-based and bandgap precision references as part of its line of data conversion products. The zener-based AD688 is a high-precision +10 and -10 V tracking reference. This device includes the basic reference cell and three additional amplifiers. The amplifiers are laser trimmed for low offset and low drift and maintain the accuracy of the reference. Low initial error and low temperature drift give the AD688 reference absolute -4-10 V accuracy performance in monolithic form.

The AD689, an 8.192 V reference, bridges the gap between 5 V and 10 V products. This device is especially useful in data conversion circuits that operate over 4-12 V but may swing over a 10% range.

The MAX2700 series (MAX2700/2701/2710) of 10 V references finds typical applications in high-resolution A/D and D/A systems and in data acquisition systems. The MAX2701 in this family is a -10 V reference.

The LTZ1000 from Linear Technology is an ultrastable reference operating at 7.2 V. This includes a heater resistor for temperature stabilization and a temperature sensing transistor, which results in very good temperature stability. Typical applications of this device are ,in voltmeters, calibrators, standard cells, scales, and low-noise RF oscillators (Linear Technology Corp., 1990).

The LT1021 is a precision reference available in three voltages: 5, 7, and 10.

These devices are intended for circuits requiring a precise 5 V or 10 V reference with very low initial tolerance.


TBL. 2 Illustrative Bandgap References

2.4.2 Bandgap References

Some illustrative bandgap references are illustrated in TBL. 2.

Typical of the lower-cost, general-purpose bandgap references is the LM136 series from National Semiconductor. The LM136 and 336 are bandgap references with an output voltage of 2.5 V and an accuracy of 1-2%. These are particularly useful in obtaining a stable reference from a 5 V logic supply. Typical applications of this series are in digital voltmeters, power supply monitors, and the like (Linear Technology Corp., 1990). The REF-03 from Precision Monolithics is a low-cost, 2.5 V bandgap reference. Silicon General's SG103 series of bandgap references is available in 13 voltage ratings, ranging from 1.8-5.6 V. The LT1019 from Linear Technology is an accurate bandgap reference, available in voltage ratings of 2.5, 4.5, 5, and 10 V. Applications for this device include A/D and D/A converters and precision regulators (Linear Technology Corp., 1990). Maxim Integrated Circuits produces a wide range of references. One such series, the MAX676/677/678 produces +4.096, 5, and 10 V calibrated, low-drift precision voltage references. One feature of the 4.096 low-dropout reference is that it operates from a 5 V -t-10% supply (Maxim Integrated Circuits, 1995). This series of references has excellent line and load regulation in addition to temperature stability. These devices find applications in high-resolution 16-bit A/D and D/A converters, precision test and measurement systems, high-accuracy transducers, and as calibrated voltage reference standards.

Micro-power voltage references, which consume as little as 10 gA operating current, are available, unlike zener-based references, which consume much larger currents. One such example is MAX872. Another micro-power reference, the MAX6120, draws a maximum current of 70 uA and operates over a 2.4-11 V input range. This is ideally suited for battery-powered systems and portable applications such as data acquisition systems (Maxim Integrated Circuits, 1996). The LM385 series of micro-power precision references operate at currents in the range of 15-20 um.


FIG. 5 Designing with voltage references: (a) A basic circuit, (b) An output voltage trimming circuit, (c) Generating a negative reference. (Maxim Integrated Circuits.)

The LT1034 micro-power precision reference from Linear Technology combines a 1.2 or 2.5 V bandgap reference with a 7 V zener-based auxiliary reference in a single package. The 1.2 V/2.5 V reference is a trimmed, bandgap voltage reference with 1% initial tolerance and guaranteed 20 ppm/°C temperature drift.

Operating on only 10 uA, the LT1034 offers guaranteed drift and good long-term stability. The 7 V reference is a subsurface zener device for less demanding applications (Linear Technology Corp., 1990). The REF1004-1.2 and REF1004-2.5 are two terminal micro-power bandgap references designed for high accuracy with outstanding temperature characteristics at low operating currents. The REF1004 is a cost-effective solution when reference voltage accuracy, low power, and long-term temperature stability are required (Burr-Brown, 1993).

2.5 Design Basics

Some basic design tips as well as some facilities available in voltage reference ICs are illustrated in FIG. 5, using the MAX873 as an example (Maxim Integrated Circuits, 1994). FIG. 5(a) shows a typical application circuit with input and output bypassing for best transient performance. FIG. 5(b) shows an output voltage trimming circuit. Although large adjustments of the output voltage may degrade its temperature performance, adjusting the output over a small range about the nominal output voltage is possible with most reference ICs.

The generation of negative reference voltages is shown in FIG. 5(c). An op amp in an inverting configuration is used, and the accuracy of the output depends on the matching of the two resistors R and R'.

3. Linear Regulators

3.1 Linear Regulator Fundamentals

FIG. 6 illustrates the basic elements of a linear regulator. The output is regulated by controlling the voltage drop across the series-pass element, a power transistor biased in the linear region. The output voltage is maintained at a constant level by changing the voltage drop across this device.

The control circuit detects the output voltage, and changes the on-resistance of the series-pass power transistor by changing its base current to keep the output voltage constant. The power dissipation in the linear regulator is a function of the difference between the input and the output voltages, output current, output driver power, and the quiescent controller power. The power dissipation in the series-pass device contributes largely to lower the efficiency of linear regulators compared to switching regulators. However, this disadvantage is insignificant in low-dropout linear regulators, which find many applications in today's sophisticated electronic and communication equipment.

A major advantage of linear regulators in comparison with switching regulators is their low noise.


FIG. 6 The basic elements of a linear regulator

3.1.1 The Series-Pass Device

The power device selected to provide the pass function must be capable of operating under very low differential input/output voltages while providing reasonable efficiency. Pass devices typically are bipolar transistors or power MOSFETs.

The first linear regulators had NPN Darlington transistors as the series-pass element. However, for low-dropout requirements, PNP transistors are more suitable, as they can maintain output regulation with very little voltage drop across it (Lee, 1989; Simpson, 1996). The dropout voltage of the linear regulator is defined as the input-output voltage differential at which the circuit ceases to regulate against further reduction in input voltage (National Semiconductor Corp., 1987). As the output requirements of the regulator grow, the gain of suitable PNP power transistors decrease, resulting in excessive base current losses. Therefore, N-channel MOSFETs are a popular choice due to their low drive current, low on- resistance and cost. Recent advances in semiconductor technology have resulted in low on-resistance P-channel devices as well. The low drive current requirement of MOSFETs reduces the quiescent current of the regulator considerably which is a major advantage of these devices. The characteristics of the series-pass device determine what the differential input/output voltage limitations are and how much quiescent power is required by the regulator. FIG. 7 shows the use of NPN Darlington and PNP transistors as the series-pass element in linear regulators. A comparison of NPN and PNP transistors with several improvements for linear regulators is found in Lee (1989).


FIG. 7 The basic linear regulator (a) With an NPN Darlington transistor and (b) With a PNP series-pass transistor

3.1.2 The Control Circuit

The control circuit samples the output voltage through a resistive divider and uses this feedback signal to control an error amplifier. Here, the regulator output is locked at a constant voltage that is a multiple of the reference voltage as determined by the voltage divider.

Control circuit characteristics directly affect system bandwidth and the achievable DC regulation. The voltage reference is used for comparison of the output voltage in the control circuit and primarily governs the steady-state accuracy of the device.

3.1.3 The Output Capacitance

The bulk capacitance maintains the output during transients. The output capacitor is required for the design to meet the specified transient requirements.

As with any control system, the voltage loop has a finite bandwidth and cannot respond instantaneously to a change in load conditions. The supply rail for many of today's microprocessors cannot vary more than 4-100 mV while handling load transients on the order of 5 A with 20-ns rise and fall times; that is, current slews at 250 A/us (Goodenough, 1996). To keep the output voltage within the specified tolerance, sufficient capacitance must be provided to source the increased load current throughout the initial portion of the transient period. During this time, charge is removed from the capacitor and its voltage decreases until the control loop can catch the error and correct for the increased current demand. The amount of capacitance used must be sufficient to keep the voltage drop within specifications. Design considerations in the selection of the capacitor value are detailed in O'Malley (1994).

3.2 Linear Regulator ICs

The linear regulator dates back to 1969. The first IC regulators, such as the LM340 or LM317, were NPN devices. Since then, many advances in technology have improved the performance of linear regulators. Regulators are available in a wide output power range. Many additional features, such as reverse-current/ overcurrent/overvoltage protection, dual mode (fixed or adjustable) operation, multiple output capabilities, thermal overload protection, and advanced control techniques, have been incorporated into linear regulator ICs since then.

Controller ICs also have been developed, which, together with external pass devices, can be used to implement linear regulators.

The basic parameters of a linear regulator are its accuracy, output current, efficiency, and the dropout voltage. Superior performance with respect to these parameters as well as low quiescent current, wide input range, and fast transient response is essential in today's applications. Special design techniques are used to develop regulators to suit particular environments such as battery-powered equipment, microprocessors, and automotive applications. National Semiconductor, Motorola, Maxim Integrated Circuits, Unitrode, Linear Technology, and Analog Devices are among the major companies producing linear regulators.

General purpose linear regulator ICs as well as those with special features such as high power output, high output current, and low-dropout voltage are available to suit a wide variety of requirements. Adjustable output as well as advanced features such as shutdown facilities to turn off all bias currents, thermal overload protection to limit the overall power dissipation in the device, and current limiting facilities are available.

3.2.1 General Purpose Linear Regulators

The LM123 is an example of a general purpose linear regulator, providing 5V at 3A, and 30 W output power (Linear Technology Corp., 1990). This and equivalent three-terminal regulators having NPN-Darlington pass transistors commonly are found in on-card regulators, laboratory supplies, and instrumentation supplies.

An example of a high-power linear regulator is the LT1038, a three-terminal, bipolar, adjustable voltage regulator capable of providing current in excess of 10 A over the 1.2-32 V range. This high-power device typically is used in battery chargers and system power supplies (Linear Technology Corp., 1990). The output voltage is adjusted by external resistors.

The LT1036 is a logic-controlled dual linear regulator, one providing 12 V at 4 A and the other 5 V at 75 mA. This device is under the control of a logic shutdown signal.

The LM137/LM337 are adjustable negative regulators, delivering up to 1.5 A of output current over an output voltage range of -1.2 V to -37 V. TBL. 3 compares these regulators.


TBL. 3 General Purpose Linear Regulators

3.2.2 Low-Dropout Linear Regulators

MAX603/604 are dual mode regulators providing either 5 V/3.3 V fixed or adjustable output. Adjustable output from 1.25-11 V may be obtained using external resistors. The P-MOSFET limits quiescent currents to as low as 35 gA. This provides several advantages over similar designs using PNP pass transistors, including longer battery life. A functional block diagram of the MAX603/604 is shown in FIG. 8(a). Its operation as an adjustable reference is shown in FIG. 8(b) (Maxim Integrated Circuits, 1996). Typical applications of these devices primarily are in battery-powered devices, pagers and cellular phones, and solar-powered instruments.

The ADP330X is a family of precision micro-power low-dropout regulators from Analog Devices. The ADP3302 contains two fully independent regulators.

Typical applications of this device are in cellular phones, note book computers, and portable instruments. MAX687 is a high-accuracy (+2%) linear regulator controller that directly drives high-gain external PNP transistors. The output current can exceed 1 A with a minimum drive current of 10 mA. It has dropout voltages of 40 mV (at 200 mA output current) and 0.8 V (at 4 A output current). An LDO controller capable of handling ultrafast current transients is the LT1575 from Linear Technology. This device, along with a discrete N-MOSFET is ideally suited for powering today's microprocessors such as the Pentium.

The nine versions of the LT1575 range from an adjustable-output controller to controllers with fixed outputs of 1.5, 2.8, 3.3, 3.5, and 5 V (Simpson, 1996). Very low-dropout voltages can be obtained, depending on the external MOSFET's on-resistance.


FIG. 8 MAX603/604: (a) Functional block diagram, (b)MAX603/604 in the adjustable mode. [Maxim Integrated Circuits, Inc.]


TBL. 4 Low-Dropout Linear Regulators

The UC3833 from Unitrode is described as a linear regulator controller suited for low-dropout, high-current regulators with a high transient response.

This device allows the use of a variety of bipolar and MOSFET power devices.

The crux of the design lies in the selection of a pass device. The design of a 3.3 V, 4 A regulator suited for today's microprocessor power supplies is described using this IC with a P-MOSFET in National Semiconductor's Linear Data Book 1 (1987).

TBL. 4 compares these regulators.

4. Switching Regulators

4.1 Switching Regulator Fundamentals

Although the linear regulator is a mature technology, due to its low efficiency and other associated disadvantages, this type of power supply tends to be unfit for most of today's compact electronic systems.

The disadvantages of the linear regulator are greatly reduced by the switching regulator. In this technology, the AC line voltage is directly rectified and filtered to produce a raw high-voltage DC. This in turn is fed into a switching element that operates at a high frequency, 20 kHz to 1 MHz, chopping the DC voltage into a square wave. The square wave then is filtered to produce a DC output. The input/output relationship of this DC/DC converter is directly related to the duty cycle of the chopping signal. Regulation is achieved by sampling the output, comparing it with a reference, and modifying the duty cycle of the chopping waveform to compensate for any drifts.

Today, most switchers operate well above 500 kHz, with new magnetics, resonant techniques, and surface mount technology extending this to several MHz.

Therefore, the associated components such as transformers and capacitors are much smaller than for linear regulators. In addition, due to the lower power loss, smaller heat sinks may be used. Therefore, the overall size of a switching regulator is smaller than an equivalent linear regulator.

The recent rapid advancement of microelectronics has created a necessity for the development of sophisticated, efficient, lightweight power supplies that have a high power-to-volume (W/in^3) ratio, with no compromise in performance.

High-frequency switching power supplies, able to meet these demands, have become the prime power source in a majority of modern electronic systems. The combination of high efficiency and relatively small magnetics results in compact, lightweight switching regulators, with power densities in excess of 100 W/in^3 versus 0.3 W/in 3 for linear regulators.

However, a major design concern in such high-frequency switching power supplies is minimization of the EMI pollution generated.


FIG. 9 The forward mode converter: (a) The basic circuit, (b) Associated waveforms

4.1.1 Modes of Operation

The DC to DC converter has two major operational modes for switching power supplies: the forward mode and the flyback mode. Although they have only subtle differences between them with respect to component arrangement, their operation is significantly different and each has advantages in certain areas of application.

Forward Mode Converters

FIG. 9(a) shows a simple forward mode converter. This type of converter can be recognized by an L-C filter section, directly after the power switch (a power transistor or power MOFSET operating between fully conducting and cutoff modes) or after the output rectifier on the secondary of a transformer.

The operation of the converter can be seen by breaking its operation into two periods: Power switch on period. When the power switch is on, the input voltage is presented to the input of the L-C section and the inductor current ramps upward linearly. During this period the inductor stores energy.

Power switch off period. When the power switch is off, the voltage at the input of the inductor flies below ground since the inductor current cannot change instantly. Then the diode becomes forward biased. This continues to conduct the current that was formally flowing through the power switch. During this period, the energy that was stored in the inductor is dumped onto the load.

The current waveform through the inductor during this period is a negative linear ramp. The voltage and current waveforms for this converter are shown in FIG. 9(b). The DC output load current value falls between the minimum and the maximum current values and is controlled by the duty cycle. In typical applications, the peak inductor current is about 150% of the load current and the minimum is about 50%. The advantages of forward mode converters are that they exhibit lower output peak-to-peak ripple voltages and they can provide high levels of output power, up to kilowatts.


FIG. 10 The flyback mode converter: (a)The basic circuit, (b) Associated waveforms


FIG. 11 Voltage and current waveforms for the discontinuous flyback mode converter

Flyback Mode Converters

In this mode of operation, the inductor is placed between the input source and the power switch, as shown in FIG. 10. This circuit also is examined in two stages: Power switch on period. During this period, a current loop including the inductor, the power switch, and the input source is formed. The inductor current is a positive ramp, and energy is stored in the inductor's core.

Power switch off period. When the power switch turns off, the inductor's voltage flies back above the input voltage, resulting in forward biasing of the diode. The inductor voltage then is clamped at the output voltage. This voltage, which is higher than the input voltage, is called the flyback voltage.

The inductor current during this period is a negative ramp.

In FIG. 10, the inductor current does not reach zero during the flyback period. This type of a flyback converter is said to operate in the continuous mode.

The core's flux is not completely emptied during the flyback period, and a residual amount of energy remains in the core at the end of the cycle. Accordingly, there may be instability problems in this mode. Therefore, the discontinuous mode is the preferred mode of operation for flyback mode converters. The voltage and current waveforms are shown in FIG. 11.

The only storage for the load in the flyback mode of operation is the output capacitor. This makes the output ripple voltage higher than in forward mode converters. The power output is lower than in forward mode converters, owing to the higher peak currents generated when the inductor voltage flies back. As they consist of the fewest number of components, they are popular in low- to medium-power applications.


FIG. 12 A typical inductor current waveform

4.1.2 A Simplified Analysis of DC/DC Converters

The input/output characteristics of all DC/DC converters can be examined by using the requirement that the initial and the final inductor currents within a cycle should be the same for steady-state operation; that is, the net energy storage within one switching cycle in each inductor should be 0. This leads to the volt-second balance for the inductor, which means that the average voltage per cycle across the inductor must be 0: that is, the volt-second products for the inductor during each switching cycle should sum to 0. This can be illustrated using a typical inductor current waveform, as shown in Figure 12.

Let the positive slope of the current be ml and the negative slope m2. For the initial and final currents to be the same, where VLon and VLoff are the voltages across the inductor during the switch on and switch off periods, respectively, and D is the duty cycle.

The output voltage of forward mode converters is given approximately by Vout "~ DVin, where D is the duty cycle of the switching waveform. Hence, this mode of operation always performs a step-down operation.

The output voltage for the flyback mode of operation is given by Vout = Vin/(1- D). Hence, flyback mode converters always are used as step-up converters.

(Cont. to part 2)

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